Direct-conversion is a wireless receiver architecture particularly suited to highly integrated, low-power terminals. Its advantage over traditional superheterodyne architectures is that the received signal is amplified and filtered at baseband rather than at some higher intermediate frequency. This architecture results in lower current consumption in the baseband circuitry and a simpler frequency plan.
In direct-conversion receivers, the most serious drawback is that the direct current (DC) offset generated by the down-conversion mixers and baseband circuitry. This offset appears in the middle of the down converted signal spectrum, corrupting the signal.
The first cause of DC offset is the transistor mismatch of the baseband components such as the down-conversion mixers and buffers. This is static DC offset. In addition, there is dynamic DC offset. One source of dynamic DC offset occurs when the local oscillator (LO) leaks into the front end of the receiver through the integrated circuit substrate. This signal is down converted to DC. Another source of dynamic DC offset occurs when the LO leaks out the antenna and reflects off external objects and back into the receiver. This too is down converted to DC.
DC offsets may be removed through capacitive coupling if the signal modulation is tolerant to the phase distortion cause by capacitor-resistor (CR) coupling. In addition, DC offsets may be estimated and digitally removed at the cost of additional hardware size and complexity.
Another problem in direct-conversion receivers is in-phase and quadrature (“IQ”) imbalance of the LO and receiver. In the art, it is well known that direct-conversion transmitter and receivers need a local oscillator with quadrature outputs for vector modulation and demodulation. However, when the quadrature outputs are not equal in amplitude and not exactly 90 degrees out of phase, demodulation becomes more difficult requiring a higher signal-to-noise ratio to properly decode the signal.
Quadrature phases are typically derived by passing a reference local oscillator through a CR-RC phase shift network. Ideally, this creates two signals with equal amplitude and 90 degrees of phase difference. However, this depends on the accuracy of resistors and capacitors which make up the phase shift network. The resistors and capacitors can vary by up to 15 percent in a typical integrated circuit causing the in-phase and quadrature components to have different amplitudes and a phase difference not equal to 90 degrees.
In addition, layout differences between the in-phase and quadrature paths can cause additional amplitude/phase imbalance. Contributing to further in-phase/quadrature imbalance is the circuits in the in-phase and quadrature paths, such as amplifiers and mixers, the physical properties of which differ slightly. Many feedback calibration schemes have been proposed and implemented to mitigate quadrature imbalance at the cost of hardware and/or system complexity.
In addition to DC offset and quadrature imbalance, radio-frequency (RF) integrated circuits suffer from self-generated interference. Specifically, signals from one part of the integrated circuit couple to another part of the integrated circuit. The RF section of an integrated circuit is the most susceptible portion since the received signal has not been fully amplified. One way to combat this problem is to turn the signal from single-ended to differential. A differential signal is comprised of a negative and a positive component. This adds to the signal's resilience to self interference.
A conventional direct-conversion receiver is illustrated in FIG. 1. As illustrated in FIG. 1, a direct-conversion receiver takes an RF signal 10 characterized by a modulation bandwidth and a center frequency. The LO produces a sinusoidal signal which has the same frequency as the RF signal center frequency, as is typical for direct-conversion receivers. As an example, a Bluetooth™ signal might be transmitted at 2440 MHz therefore the LO may produce a 2440 MHz sinusoidal signal for down conversion.
Furthermore, the receiver multiplies the RF signal not with one but with two different phases 11, 12 of the LO. The two phases 11, 12 of the local oscillator are 90 degrees apart and thus, are known as the in-phase (I) 11 and quadrature (Q) 12 components. Through this disclosure, the in-phase local oscillator signal is denoted LOI and the quadrature local oscillator signal is denoted LOQ. The mixer outputs 13, 14 are known as baseband signals since they are at a lower frequency than the RF signal. The baseband signals are in-phase and quadrature corresponding to the in-phase and quadrature local oscillator signals. The baseband signals are low pass filtered as to remove unwanted interfering signals. Through this disclosure, the in-phase baseband signal is denoted BBI and the quadrature baseband signal is denoted BBQ. The resulting filtered baseband signals 15, 16 can be represented by Equations 1 and 2.BBI=RF×LOI  Equation 1BBQ=RF×LOQ  Equation 2
Another conventional direct-conversion architecture is shown in FIG. 2. This differential direct-conversion architecture is more resilient to self-generated noise than the one illustrated in FIG. 1. In FIG. 2, the RF input signal 200 is converted by a balun 220 to a differential signal composed of positive and negative components 201, 202 respectively. The relationship between the RF input 200 and the differential components 201, 202 are described by Equation 3.RF=(RFpos−RFneg)  Equation 3
Similarly, the differential direct-conversion architecture shown in FIG. 2 uses differential LO signals to mix the RF signal down to baseband. The polyphase network 205 is a circuit which converts the local oscillator's voltage waveform 203 into four voltage waveforms 206, 207, 208, 209 at the same frequency as the LO 203 but at 0, 180, 90, 270 degrees offset compared to the LO signal 203 respectively.
Collectively, these four signals 206, 207, 208, 209 are referred to as polyphase local oscillator signals. To facilitate the description of this embodiment, these signals are denoted 206, 207, 208, 209 as LO0, LO180, LO90, LO270 corresponding to their phase shift compared to the local oscillator 203. It is well known in the art that shifting a sinusoidal signal 180 degrees in phase is the same as inverting the signal. Therefore, the equivalent single-ended in-phase and quadrature LO signals are described mathematically as in Equations 4 and 5.LOI=LO0−LO180  Equation 4LOQ=LO90−LO270  Equation 5
The differential RF signal 201, 202 is then routed to the differential mixers 210, 211 where it is multiplied by the differential local oscillator signals. At the first mixer 210, the differential RF signal is multiplied by the in-phase LO (LOI) to generate the differential in-phase baseband signal 212, 213 (BBI). Likewise, at the second mixer 211, the differential RF signal is multiplied by the quadrature LO (LOQ) to generate the differential quadrature baseband signal 214, 215 (BBQ). Equations 6 and 7 describe the mixing process of the differential signals to generate the BBI and the BBQ.BBI=(BBI,pos−BBI,neg)=(RFpos−RFneg)×(LO0−LO180)  Equation 6BBQ=(BBQ,pos−BBQ,neg)=(RFpos−RFneg)×(LO90−LO270)  Equation 7
As in the single-ended case, the baseband signals 212, 213, 214, 215 can be filtered to remove unwanted interfering signals to produce filtered baseband signals 216, 217, 218, 219.
Now, to elucidate the problems with direct-conversion receivers, DC offset and imbalance distortions will be added to Equations 6 and 7. DC offsets are added to the output of the mixers. DC 1 represents the differential DC offset of the first mixer 210 and DC2 represents the differential DC offset of the second mixer 211. Likewise the amplitude and phase imbalance of the mixers and the polyphase LO signals can be accounted for at the output of each mixer. A complex multiplicative term, A1ejP1, represents a random amplitude variation (A1) and a random phase variation (P1) introduced by the first mixer 210 and the signal path and LO path connected to the mixer. Likewise, A2eP2 represents a random amplitude and phase variation introduced by the second mixer 211 and the signal and LO paths connected thereto. Thus, with these distortions added, Equations 6 and 7 become Equations 11 and 12.BBI=(RFpos−RFneg)×(LO0−LO180)×A1ejP1+DC1  Equation 11BBQ=(RFpos−RFneg)×(LO90−LO270)×A2ejP2+DC2  Equation 12
As seen in Equations 11 and 12, the baseband in-phase and quadrature signals imbalance grows as A1 and A2 differ and as P1 and P2 differ. As the imbalance increases, it is harder for the signal to be received and decoded. Likewise, as DC1 and DC2 get larger, and thus depart from the ideal of no DC offset, it becomes more difficult for the signal to be received and decoded.